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  A Dual Polarized Dual Band Microstrip Antenna for Wireless Communications Bjom Lindmark Allgon System AB Box 541,183 24 Taby, Sweden phone: +46-8-540 82639 e-mail: Abstract-The demand for antennas for mobile wireless applications has increased dramatically over the last 10 years. Today we have a number of land and satellite based systems for wireless communications using a wide range of frequency bands. Not only do we see an increase in the number of subscribers in the different systems but also a demand for dual or multi band equipment capable of han- dling two or more frequency bands. Our paper describes a dual polarized, dual band antenna element suited for use in a base station antenna. The antenna element is an aperture coupled, stacked patch, maintaining the symmetry needed for dual polarization operation. The total height is less than 0.15 h at the lowest frequency and the element provides broadside radiation. Whereas most previously presented dual band elements are linearly polarized microstrip anten- nas which utilize a number of narrow-band resonators, our design has all the broad-band and dual-polarized character- istics of traditional aperture coupled patch antennas. The bandwidth for Return Loss > 10 dB of the element covers both the 880-960MHz GSM (Global System for Mobile Communication) band and the 17 10- 1880 MHz DCS (Digital Cellular System) frequency band. The measured isolation between the ports corresponding to the two or- thogonal polarizations is greater than 32 dB in both bands. TABLE F CONTENTS INTRODUCTION DUAL BAND LEMENT MEASURED ESULTS ND DISCUSSION CONCLUSIONS NDFUTURE ORK REFERENCES BIOGRAPHY 1. INTRODUCTION Due to the capacity problems today encountered in the AMPS 800 MHz and GSM 900 MHz wireless communica- tion systems, many operators have acquired licenses in the 1800 or 1900 MHz bands. Since a major problem during the deployment of a cellular radio network is to find suitable Department of Microwave Technology Chalmers IJniversity of Technology 412 96 Gothenburg, Sweden e-mail: sites for the base stations, one can expect these operators to use their existing sites for the new 1800 or 1900 MHz base station wherever possible. Then, one 'possibility is to re- place an existing GSM or AMPS antenna with a dual band GSM/DCS or AMPSPCS antenna. In this paper we present an antenna element suited for the GSM/DCS bands, i.e. 880-960 MHz and 1710-1880 MHz. A dual band base sta- tion antenna would have a linear array of such elements po- sitioned along the vertical axis. In order for the dual band antenna to provide a clear advan- tage to the operator, it should have only one input for both bands. This allows the use of only one feeder cable thereby reducing the cost of both equipment acid installation. The cnly extra equipment ineeded is a bandl-separating filter at the base station, and such filters are fairly uncomplicated. The elements in such a base station antenna should therefore have full dual band capability with respect to VSWR and radiation properties. Recent years haw seen an increased interest in using polarization diversity at the base station; e.g. [ 1, 21, and therefore we seek an element also capable of dual polarization operation. Much work has been done on extending the use of micro- strip patch antennas to dual band or multi-frequency opera- tion. This includes the use of multiple aperture coupled resonators [3], the use of slots in the patch [4], the separa- tion of the bands into two channels feelding different layers of a patch structure [5] and the use of stacked patches [6]. An approach using printed dipoles for dual band operation was presented in [7]. All the above work regards linearly or circularly polarized elements. Numerous authors have pre- viously investigated dual polarized microstrip patch ele- ments, e.g. [8-111. In the present work we extend the use of microstrip patches to dual band and duad polarization opera- tion. 2. DUAL AND ELEMENT Geometry and electrical operation The suggested dual band element consists of a three-layer aperture coupled patch structure as shown in figures 1-2. The three patches are centered over a cross-shaped aperture 333 0-7803-43 1 1 5/98/ 10.00 1998 IEEE  in a 2 mm thick and 250 mm wide Aluminum reflector. The last dimension of this reflector, i.e. its length, is assumed to be sufficiently long not to have any effect on the antenna characteristics discussed in this paper. In the base station antenna application for this element the length-wise direc- tion would be the vertical direction and the plane orthogonal to this direction is thus from hereon referred to as the hori- zontal plane. The patches are separated by Rohacell foam with a permittivity of approximately 1.05. The cross-shaped aperture is excited by a microstrip feed network etched on a 0.762 mm thick DiClad 527 teflon substrate with permittiv- ity 2.55. The feed is divided into two parts, one part excites the vertically polarized channel and one excites the hori- zontally polarized channel. These two channels are referred to as channel 1 and 2 respectively. The feed consists of a 50 R ine which divides into two 50 L2 lines. These two lines excite the aperture in a symmetrical fashion so that for ex- ample the feed of channel 1 excites a voltage over the verti- cally aligned parts of the cross-shaped aperture. The lines end in open circuit stubs. In order to match the input im- pedance to 50 2 in both frequency bands a small amount of symmetrical capacitive tuning was applied on both channels. Figure 2 shows these short sections of wider microstripline some 30 mm before the aperture. In the lower band the antenna radiates through the same mechanism as in any aperture coupled patch element: The aperture couples power to the large radiating patch and the patch and the aperture both radiate into the far-field. In or- der to provide dual polarization operation a cross-shaped aperture is used with a feed network similar to the feed de- scribed in [8]. This feed arrangement provides the symme- try necessary for high port-to-port isolation and good polari- zation purity. Since the feed of both polarization channels are positioned in the same layer it is necessary to have mi- crostrip lines crossing each other at some point. Thus the location of an airbridge is indicated in figure 2. The size and position of the large patch is chosen for good performance in the lower frequency band. The dimensions are very similar to a linearly polarized single band aperture coupled patch antenna. The large patch has another cross- shaped aperture which allows for coupling of upper band frequencies to a patch positioned on top of it. This aperture was found to lower the resonance frequency of the patch a few percent and the patch size was therefore adjusted ac- cordingly. The patch below the largest patch couples the power in the upper band from the large slot in the ground- plane to the small slot in the large patch. The initial designs intended for single linear polarization included a small slot in the groundplane for the upper frequency band, but we found that dual band operation was possible with the large slot alone. The position and sizes of the two small patches are chosen for good performance in the upper frequency band. We can expect the large patch to function as a groundplane for the top patch. This hypothesis is supported by the simi- lar radiation patterns found in the horizontal plane for the two bands and the fact that the ratio between the size of the largest and the reflector width is approximately equal to ratio of the top patch and the large patch. In the case of a linearly polarized element, the width of the patches would provide some degrees of freedom to achieve a desired beamwidth, especially in the upper band. The direct back radiation from the aperture is reduced to practically zero by the use of a shielding conducting box as described in [12]. In our case the dimensions of the box shown in figure2 are lOOx 1OOx 8 mm and it is made of Aluminum and filled with air. The box is electrically grounded via 12 screws symmetrically placed on the 4 sides extending to the 250 wide reflector. Radiating patch (50 mm) for DCS - coupling slot for Patch (60 mm) / DCS-band for DCS Microstrip feedline on 0.762 mm DiClad 527 _I Shielding cage substrate grounded at 12 points Cross-section of antenna element igure I a: 334  /' 250 rr reflector Fisre 6: Side view of the antenna element. Foam layers, eed tzetwork and shielding cage not shown. The antenna element is centered on a 6.50 mm on and 2.50 mm wide rejlector. Reflector: 250 mm wide A C 54 x 5 mm \ Shieldinn Pond 1 YUWF _. .., for back radiation 100 100 mm aperture in the groundplane _ Dc patd Feed nemrk of port 1 LlGS coupling patch: 60 x 60 mm Vertical direction Figure 2: Top view of antenna element showing parches afldfeed network 3. MEASURED SULTS AND DISCUSSION Return Loss andlsolation The S-parameters of he antenna element described above were measured using a HP 8753D etwork analyzer. The result is shown in figure 3 where SI2 has been omitted since it is identical to 521. The return loss in the iower band (GSM) is greater than 15 dB for both channels. In the upper band DCS) he return loss is greater than 10 dB. The bandwidths for return loss > lOdB are 14.3 around 920 M z and 14.7 around 795 MHz for channel 1. It seams more difficult to achieve a high return loss design in the up- per band than in the lower with this design. Although the electrical dimensions in this case are twice as large in the upper band, the problem of obtaining sufficient coupling from the large aperture in the groundplane via the coupling patch to the aperture in the large patch so far limits the per formance of the element. The port-to-port isolation is greater than 32dB in both bands as seen from the S21-parameter in figure 3. However, in the region between the frequency bands of interest we see two points of resonance where the coupling is much larger. From measurements on the feed network alone, i.e. with no aperture in the groundplane present, we coufd conclude that the resonance phenomenon was related to the exact configu- ration of the feed network. It is our belief that the coupling primarily occurs where the open circuit stubs run parallel to a feedline of the other polarization. In this case we chose the length of the open circuit stubs so that the no resonance would be present in the frequency bands of interest. 335    -5 -1 -1 5 (I) 0 c rn 20 . -25 -30 -35 -40 800 1000 1200 1400 1600 1800 2000 frequency (MHz) Figure 3: Measured return loss for the two channels. SI olid S22 dashed and S21 dotted. The GSMfiequency band 880-960 MHz) and DCSfiequency bands I 710-1880 MHz) are indicated by dotted vertical lines. Radiation Patterns The far-field radiation pattern of the antenna was measured in an indoor anechoic chamber. The antenna element was aligned as shown in figure 2. It was mounted on a 250 mm wide and 650 mm long reflector. Since we are primarily interested in the horizontal pattern of a linear array of the element, the reflector was loaded with absorbing material at the top and bottom edges. This minimized the E-plane dif- fraction from the vertically polarized channel 1. Figures 4 (a-b) show the horizontal plane patterns at 920 MHz and 1795 MHz. The power is normalized to the co-polar level of channel 1 At both frequencies we have broadside radiation. The H-plane beamwidth of channel 1 is 72.1 and 74.4 at these two frequencies. The correspond- ing E-plane beamwidth of channel 2 is 61.0 and 54.1'. Since the antenna patch structure as well as the feed is es- sentially symmetric with respect to the vertical axis, we ex- pect low cross-polarization in the horizontal plane. How- ever, we see a -25 dB cross-polar level around boresight at 920MHz and 1795 MHz and this indicates that some asymmetry is indeed present. Although the isolation at these frequencies is in excess of 30dB, we do not rule out the possibility that the same coupling causing the resonance phenomenon discussed previously also causes an unbalance in the excitation of the cross-shaped aperture in the ground- plane 336


Jul 23, 2017
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